Method for controlling resonant power conversion device, and resonant power conversion device

ABSTRACT

A method for controlling a resonance type power converter including a first resonance circuit (L0, C0) and a shunt circuit (3), which converts and outputs the power of the DC power supply, shunting a current flowing into a first capacitor (CS) by controlling a second switching element (S2) during a predetermined period within turn-off period of a first switching element (S1), the first capacitor connected in parallel to the first switching element (S1), the second switching element (S2) included in the shunt circuit (3), and the first switching element (S1) operated in response to the resonance of the first resonance circuit (L0, C0).

TECHNICAL FIELD

The present invention relates to a method of controlling a resonance type power converter and a resonance type power converter.

BACKGROUND ART

In order to reduce the applied voltage to switching element constituting a class E circuit, a class E/F circuit comprising the class E circuit, and a resonance circuit for superimposing harmonics of an integer multiple of the operating frequency during soft switching operation is known with (Patent Document 1: JP 2011-101408 A).

PRIOR ART DOCUMENT Patent Document

[Patent Document 1] JP 2011-101408 A

SUMMARY OF INVENTION Problems to be Solved by Invention

In the prior art, there is a problem that since the harmonics of the integer multiple of the operating frequency are superimposed by the resonance circuit, the switching loss is increased as compared with a resonance type power converter using only the class E circuit.

An problem to be solved by the present invention is to provide a method for controlling a resonance type power converter and a resonance type power converter that can reduce the switching loss generated during the soft switching operation.

Means for Solving Problems

The present invention solves the above problems by shutting a current flowing into a first capacitor connected in parallel to a first switching element by controlling a second switching element included in a shunt circuit, during a predetermined period within turn-off period of the first switching element which operates in response to resonance of the first resonance circuit.

Effect of Invention

According to the present invention, it is possible to reduce the switching loss generated during the soft switching operation.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a configuration diagram of a power conversion system comprising a resonance type power converter according to a first embodiment;

FIG. 2 is a diagram for explaining a control signal to a second switching element according to the first embodiment.

FIG. 3 is an exemplary operation of the resonance type power converter according to the first embodiment.

FIG. 4 is an input impedance characteristics of the resonance type power converter according to the first embodiment.

FIG. 5 is a configuration diagram of a power conversion system comprising a resonance type power converter according to a reference example.

FIG. 6 is an exemplary operation of the resonance type power converter shown in FIG. 5.

FIG. 7 is an exemplary operation of a resonance type power converter according to a comparative example.

FIG. 8 is an example of a comparison result of the resonance type power converter according to the first embodiment, the resonance type power converter according to the reference example described with reference to FIGS. 5 and 6, and the resonance type power converter according to the comparative example described with reference to FIG. 7.

FIG. 9 is a configuration diagram of a power conversion system comprising a resonance type power converter according to a second embodiment.

FIG. 10 is a diagram for explaining a control signal to a second switching element according to the second embodiment.

FIG. 11 is an exemplary operation of the resonance type power converter according to the second embodiment.

FIG. 12 is the input impedance characteristics of the resonance type power converter according to the second embodiment.

MODE(S) FOR CARRYING OUT THE INVENTION

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

First Embodiment

FIG. 1 is a diagram illustrating a configuration of a power conversion system comprising a resonance type power converter according to a first embodiment. A power conversion system 10 according to the first embodiment is utilized in a hybrid vehicle capable of traveling a motor as a driving source and an electric vehicle for traveling a motor as a driving source, or the like. Note that there is no particular limitation on a situation in which the power conversion system 10 is used.

As shown in FIG. 1, the power conversion system 10 is a system in which the DC power of an input voltage source 1 is converted and supplied to the load 2 by the resonance type power converter according to the present embodiment. The resonance type power converter of the present embodiment is an inverter, and controls output power according to the state of supplying power to the load 2.

The input voltage source 1 generates and outputs DC power. For example, in the input voltage source 1, an AC voltage input from the commercial power supply (e.g., 200V) is rectified by a rectifier circuit (not shown), and smoothed by a smoothing circuit (not shown), so that the AC voltage is converted to a DC voltage. Then, DC-DC converter (not shown) converts to a DC voltage of a predetermined target voltage. Incidentally, the configuration of the input voltage source 1 is not limited, it may be those for outputting a predetermined DC voltage.

A inverter converts the DC voltage input from the input source 1 into an AC voltage. The inverter comprises an input coil L_(c) a first switching element S₁, a shunt capacitor C_(s), a series resonance circuit configured with a coil L₀ and a capacitor C₀, and a shunt circuit 3. The inverter is a so-called class E inverter capable of class E operation.

The input coil L_(c) is connected series between a high-potential output terminal of the input voltage source 1 and the first switching element S₁. Specifically, one end of the input coil L_(c) is connected to the input voltage source 1, and the other end of the input coil L_(c) is connected to the first switching element S₁. The input coil L_(c) supplies a constant current to the first switching element S₁, when the input voltage source 1 outputs a constant DC voltage. The input coil L_(c) is a so-called choke coil.

The first switching element S₁ switches between an on-state and an off-state in accordance with a control signal input from the controller 4. The first switching element S₁ switches from the off-state to the on-state, when the first switching element S₁ is turned on. Conversely, the first switching element S₁ switches from the on-state to the off state, when the first switching element S₁ is turned off.

The first switching element S₁ will be described as an MOSFET (Metal Oxide Semiconductor Field Effect Transistor) in the present embodiment, but the first switching element S₁ is not specifically limited. The first switching element S₁ may be, for example, a bipolar transistor capable of being controlled by a current, an IGBT (Insulated Gate Bipolar Transistor) capable of being controlled by a voltage, or the like. A drain terminal of the first switching element S₁ is connected to the other end of the input coil L_(c). A source terminal of the first switching element S₁ is connected to a low-potential side output terminal of the input voltage source 1.

The control signal is input from the controller 4 to a gate terminal of the first switching element S₁. The control signal is a pulse signal repeating high-level and low-level at a predetermined frequency. The predetermined frequency is a frequency at which the first switching element S₁ is switched, in the following, also referred to as a switching frequency f_(s). When the control signal changes from high-level to low-level, the first switching element S₁ is turned off, and when the control signal changes from a low-level to high-level, the first switching element S₁ is turned on.

The first switching element S₁ has a diode inside the device. As shown in FIG. 1, an anode terminal of the diode is connected to the source terminal of the first switching element S₁, and a cathode terminal of the diode is connected to the drain terminal of the first switching element S₁. Such a diode can protect the first switching element S₁ from inverse electromotive force generated by the first switching element S₁ switching.

The shunt capacitor C_(s) is connected to the first switching element S₁ in parallel. A current flow int to the shunt capacitor C_(s) during the off-state of the first switching element S₁. The shunt capacitor C_(s) is a capacitor for storing electric energies. After a voltage across the shunt capacitor C_(s) reach peaks, the shunt capacitor discharges to release electric energies. When the voltage across the shunt capacitor C_(s) becomes zero, the first switching element S₁ is switched from the off-state to the on-state. The first switching element S₁ control by the controller 4 will be described later.

The coil L₀ and capacitor C₀ are designed such that the resonance frequency matches the switching frequency f_(s), and the coil L₀ and capacitor C₀ comprise a series resonance circuit. One end of the coil L₀ is connected to the drain terminal of the first switching element S₁ and to the other end of the input coil L_(c). The other end of the coil L₀ is connected to one end of the capacitor C₀. The other end of the capacitor C₀ is connected to the load 2, which will be described later. In the following, for convenience of explanation, the series resonance circuit configured with the coil L₀ and the capacitor C₀ will be referred to as a first resonance circuit. Due to the resonance characteristics of the first resonance circuit, the output voltage of the inverter is a voltage that oscillates at resonance frequency (the switching frequency f_(s)) along a sine wave.

The load 2 is provided on the output side of the inverter. The load 2 may be, for example, a motor, a battery comprised of a secondary battery, or the like. If the load 2 is a battery, the battery is connected to the inverter via a rectifier. In the present embodiment, the type of the load 2 is not particularly limited.

The shunt circuit 3 is connected in parallel to the shunt capacitor C_(s) to shunt the current flowing into the shunt capacitor C_(s). In the present embodiment, the shunt circuit 3 is configured with the series resonance circuit of a coil L and a capacitor C₁ and a second switching element S₂ connected in series to the series resonance circuit. One end of the coil L is connected to the other end of the input coil L_(c), and the other end of the coil L is connected to one end of the capacitor C₁. In the following, for convenience of explanation, the series resonance circuit configured with the coil L and the capacitor C₁ will be described as a second resonance circuit. Incidentally, shunting the current flowing into the shunt capacitor C_(s) means that it makes the current to flow separately from the main current, when the current flowing into the shunt capacitor C_(s) is regarded as the main current. In the following description, for convenience of explanation, a shunt current includes the current itself of the main current flowing in another direction without being divided from the main current.

The second switching element S₂ switches between an on-state and an off-state in accordance with a control signal input from the controller 4. The second switching element S₂ switches from the off-state to the on-state, when the second switching element S₂ is turned on. Conversely, the second switching element S₂ switches from the on-state to the off-state, the second switching element S₂ is turned off.

In this embodiment, the second switching element S₂ will be described as a MOSFET, like the first switching element S₁, but second switching element S₂ is not specifically limited. The second switching element S₂ may be, for example, a bipolar transistor capable of being controlled by a current, an IGBT (Insulated Gate Bipolar Transistor) capable of being controlled by a voltage, or the like. A drain terminal of the second switching element S₂ is connected to the other end of the coil L. A source terminal of the second switching element S₂ is connected to a low-potential side output terminal of the input voltage source 1.

In this embodiment, the second switching element S₂ is arranged so that potential of the source terminal of the second switching element S₂ is the same as potential of the source terminal of the first switching element S₁. This reduces influence of noise on the second switching element S₂. The arrangement relation between the second switching element S₂ and the second resonance circuit is not particularly limited. For example, unlike FIG. 1, the second switching element S₂ may be arranged at high potential side of the input voltage source 1 than the second resonance circuit.

The control signal is input from the controller 4 to a gate terminal of the second switching element S₂. The control signal becomes high-level at a predetermined timing in accordance with the state of the first switching S₁. Thereafter, the control signal becomes low-level after a predetermined period has elapsed in accordance with the state of the first switching element S₁. When the control signal changes from high-level to low-level, the second switching element S₂ is turned off, and when the control signal changes from low-level to high-level, the second switching element S₂ is turned on. The control signal is switched to high-level or low-level in accordance with timing the first switching element S₁ is turned off or turned on. The second switching element S₂ control by the controller 4 will be described later.

The second switching element S₂ has a diode inside the device. As shown in FIG. 1, an anode terminal of the diode is connected to the source terminal of the second switching element S₂, and a cathode terminal of the diode is connected to the drain terminal of the second switching element S₂.

The controller 4 is configured with a microcomputer having a CPU (Central Processing Unit), a ROM (Read Only Memory), and a RAM (Random Access Memory), or a FPGA (Field Programmable Gate Array).

The controller 4 controls output power of the inverter by controlling the first switching element S₁ of the inverter. In addition, the controller 4 controls the second switching element S₂ to shunt the current flowing into the shunt capacitor C_(s) and causes the current to flow into the shunt circuit 3.

First, the control of the first switching element S₁ by the controller 4 will be described. The controller 4 generates a control signal for turning on and off the first switching element S₁, and outputs it to the gate terminal of the first switching element S₁. For example, the controller 4 generates a pulse signal having the switching frequency f_(s) based on a reference clock. The controller 4 amplifies the pulse signal to a level at which the first switching element S₁ can be driven by a drive circuit (not shown) and outputs the amplified pulse signal as a control signal to the gate terminal of the first switching element S₁. This causes the first switching element S₁ to repeat turn-on or turn-off.

In addition, to control the first switching element S₁, a detection value is input to the controller 4 from a current sensor (not shown) for detecting the output current of the first switching element S₁. The controller 4 performs feedback control based on the output current value. For example, the controller 4 changes or adjusts frequency and duty ratio of the control signal based on the output current value. As a result, the switching frequency f_(s) and the duty ratio of the first switching element S₁ are changed, and the output power of the inverter can be controlled. Incidentally, the duty ratio is a value indicating the ratio of the on-period to a unit period when total period of on-period and off-period of the first switching element S₁ is the unit period.

In addition, the controller 4 generates a control signal for turning on the first switching element S₁ when the voltage applied across the drain terminal and the source terminal of the first switching element S₁ is zero. Generally, in first switching element S₁, an on-resistance exists between the drain terminal and the source terminal due to the inner structures. For example, when a switching element is turned on while a predetermined voltage is applied across both terminals of the switching element, power consumption based on the voltage across both terminals and the on-resistance of the switching element is generated, thereby lowering the power conversion efficiency of the inverter (also referred to as switching loss).

In the inverter having the first resonance circuit configured with the coil L₀ and the capacitor C₀ as in the present embodiment, the resonant characteristics of resonance circuit cause the voltage applied across the drain terminal and the source terminal of the first switching element S₁ to be a voltage that varies with time along a sine wave. Therefore, for example, when the first switching element S₁ is turned on while the voltage across the drain terminal and the source terminal is zero voltage, the power consumed by the first switching element S₁ is significantly reduced, and the power conversion efficiency of the inverter can be improved. In the following description, for convenience of explanation, the operation of such the first switching element S₁ is referred to as ZVS (Zero Voltage Switching), zero-voltage switching, or soft-switching. The operation of ZVS or the like includes an operation in which the first switching element S₁ is turned off when the voltage across the drain terminal and the source terminal is zero voltage.

Next, the control of the second switching element S₂ by the controller 4 will be described. The controller 4 generates a control signal for turning on and off the second switching element S₂, and outputs it to the gate terminal of the second switching element S₂. When generating the control signal of the second switching element S₂, the controller 4 amplifies the control signal to a level at which the second switching element S₂ can be driven by the drive circuit, like the first switching element S₁ control.

The timing to turn on or off the second switching element S₂ and the turn-on period of the second switching element S₂ will be described with reference to FIG. 2.

FIG. 2 is a diagram for explaining the control signal to the second switching element S₂. In FIG. 2, the upper graph shows the control signal to the first switching element S₁ and the lower graph shows the control signal to the second switching element S₂.

As shown in FIG. 2, the controller 4 controls the second switching element S₂ while the control signal to the first switching element S₁ is low-level. The controller 4 turns on the second switching element S₂ for a predetermined period within turn-off period of the first switching element S₁. The controller 4 then turns off the second switching element S₂ after a predetermined time. Specifically, the controller 4 turns on the second switching element S₂ later than the first switching element S₁ is turned off. When the second switching element S₂ is on, the controller 4 turns off the second switching element S₂ earlier than the first switching element S₁ is turned on. This allows current flowing into the shunt capacitor C_(s) to be shunted during turn-off period of the first switching element S₁.

In the example of FIG. 2, the controller 4 turns off the first switching element S₁ (time T₁) and turns on the second switching element S₂ (time T_(s2_ons)) after a predetermined time has elapsed. The controller 4 then turns on the second switching element S₂ during the turn-on period T_(s2_on), and then turns off the second switching element S₂ (time Ts2_one). When control to the second switching element S₂ is completed, the controller 4 turns on the first switching element S₁ (time T₂).

Next, with reference to FIGS. 1 and 3, the operation of the resonance type power converter according to the present embodiment will be described. FIG. 3 is an exemplary operation of the resonance type power converter according to the present embodiment. FIG. 3 (a) shows the control signal to the first switching element S₁ and corresponds to the upper plot of FIG. 2. FIG. 3 (b) shows the control signal to the second switching element S₂ and corresponds to the lower graph of FIG. 2. FIG. 3(c) shows the voltage V_(s1_ds) applied across the drain terminal and the source terminal of the first switching element S₁, and FIG. 3(d) shows the current I_(cs) flowing into the shunt capacitor C_(s). FIG. 3(d) shows the current I_(c) flowing into the shunt circuit 3. The operation of the resonance type power converter will be described in the order of the time series shown in FIG. 3.

The operation of the resonance type power converter from the turn-off of the first switching element S₁ to the turn-on of the second switching element S₂ will be described. As shown in FIGS. 3(b), (d), (e), when the first switching element S₁ is turned off by the controller 4, the voltage V_(s1_ds) rises from the zero voltage and the current Is rapidly rises from the zero current. This indicates that when the first switching element S₁ is turned off, the charging of the shunt capacitor C_(s) begins, and the voltage applied across both terminals of the first switching element S₁ connected in parallel to the shunt capacitor C_(s) begins to rise. Once the first switching element S₁ is off, a current corresponding the capacitance value continues to flow into the shunt capacitor C_(s) until the shunt capacitor C_(s) is fully charged. Thus, the voltage V_(s1_ds) increases until the shunt capacitor C_(s) is fully charged.

Next, the operation of the resonance type power converter when the second switching element S₂ is turned on will be described. As shown in FIGS. 3(b), (d), (e), when the second switching element S₂ is turned on by the controller 4 with the first switching element S₁ off, the current I_(c) rises from zero current and the current I_(cs) rapidly rises. This indicates that, as shown in FIG. 1, by turning on the second switching element S₂, the second resonance circuit conducts, and the current shunted from the ICs (hereafter simply referred to as shunt current) begins to flow into the capacitor C₁. The current flow into the shunt circuit 3 significantly reduces the current I_(cs) flowing into the shunt capacitor C_(s).

Next, the operation of the resonance type power converter during the turn-on period of the second switching element S₂ will be described. As shown in FIG. 3(c), the rising of the voltage V_(s1_ds) is suppressed. This indicates that during the turn-on periods of the second switching element S₂, the voltage applied across both terminals of the shunt capacitor C_(s) is suppressed because the current planned to flow into the shunt capacitor C_(s) is shunted. As shown in FIG. 3(e), the current I_(c) is a current represented by a sine wave. In this embodiment, as shown in FIG. 1, the shunt circuit 3 includes the second resonance circuit. Therefore, the second resonance circuit resonates during the turn-on of the second switching element S₂. The resonant characteristics of the second resonance circuit will be described later.

Next, the operation of the resonance type power converter from the turn-off of the switching element S₂ to the turn-on of the switching element S₁ will be described. As shown in FIG. 3(e), when the second switching element S₂ is turned off, the current I_(c) indicates a zero current. This indicates that the second switching element S₂ is turned off and no current flows into the shunt circuit 3. Further, as shown in FIGS. 3 (c), (d), the voltage V_(s1_ds) shows a shape such that toward the zero voltage along the sine wave, and the current ICs shows a negative current. During the turn-on period of the second switching element S₂, resonance occurred by the coil L₀ and the capacitor C₀ causes to start discharging at the shunt capacitor C_(s). Thus, in a period from the time the second switching element S₂ is turned off to the time the first switching element S₁ is turned on, the shunt capacitor C_(s) discharges, and the voltage applied across both terminals of the first switching element S₁ decreases toward zero voltage.

As described above, in the resonance type power converter according to the present embodiment, by controlling the second switching element S₂ during turn-off period of the first switching element S₁, the current flowing into the shunt capacitor C_(s) can be divided to suppress the voltage applied across both terminals of the first switching element S₁. As a result, the peak voltage value can be reduced, and the load on the first switching element S₁ can be reduced.

Next, the resonant characteristics of the second resonance circuit comprising the shunt circuit 3 will be described. The controller 4 sets turn-on period T_(s2_on) of the second switching element S₂ for turn-off period T_(s1_on) of the first switching element S₁ to satisfy Equation (1) below.

[Number 1]

T_(s2_on)<T_(s1_on)  (1)

Further, the controller 4, in relation to the resonance frequency f_(s1) defined by the coil L and the capacitor C₁, to satisfy Equation (2) below, sets the turn-on period T_(s2_on) of the second switching element S₂.

$\begin{matrix} {\left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack \mspace{616mu}} & \; \\ {T_{s\; 2\_ \; {on}} = {{N \times T_{s1}} = {D_{s2} \times T_{s\; 1}}}} & (2) \\ {\left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack \mspace{616mu}} & \; \\ {T_{s1} = {\frac{1}{f_{s1}} = {2\pi \sqrt{LC}}}} & (3) \end{matrix}$

However, T_(s2_on) is the turn-on period of the second switching element S₂, N is a real number, T_(s1), as shown in Equation (3), the reciprocal of the resonance frequency f_(s1) defined by the coil L and the capacitor C₁, D_(s2) is a ratio (duty ratio) indicating the on-period of the second switching element S₂ per unit time.

In the present embodiment, the controller 4 sets the turn-on period T_(s2_on) of the second switching element S₂ so as to satisfy the above Equations (1) to (3). In other words, the controllers 4 control the duty ratio to the second switching element S₂. As a result, while a shunt current, which is a resonant current, can be periodically flown to the shunt circuit 3, and the shunt current can be flown to the shunt circuit 3 at a predetermined time within turn-off period of the first switching element S₁. In addition, during the turn-on of the first switching element S₁, no current flows into the shunt circuit 3, so that the shunt can be performed intermittently.

Next, with reference to FIG. 4, the input impedance characteristics of the resonance type power converter according to the present embodiment will be described. FIG. 4 is the input impedance characteristics of the resonance type power converter according to the present embodiment. FIG. 4 is the input impedance characteristic when viewing the load 2 side from the input voltage source 1 among the resonance type power converter shown in FIG. 1.

The operating band of the first switching element S₁ in FIG. 4 is a frequency band for executing a soft switching operation. The input impedance corresponding to this operating band is the impedance characteristic which becomes inductive for executing the soft switching operation. The controller 4 performs the first switching element S₁ turn-on control and turn-off control in the operating band of the first switching element S₁.

The sub-resonance point shown in FIG. 4 is the resonance frequency f_(s1) defined by the coil L and the capacitor C₁. The controller 4 according to the present embodiment controls the second switching element S₂ so that the resonance frequency f_(s1) is in a higher frequency band than the operating band of the first switching element S₁. Thus, while suppressing the influence on the soft switching operation performed by the first switching element S₁, it is possible to shunt the current flowing into the shunt capacitor C_(s).

Next, the operational advantage of the resonance type power converter according to the present embodiment will be described with reference to a reference example and a comparative example.

FIG. 5 is a configuration diagram of a power conversion system comprising a resonance type power converter according to the reference example. FIG. 6 shows an example of the operation of the resonance type power converter shown in FIG. 5. Further, FIG. 7 shows an example of the operation of the resonance type power converter according to the comparative example. Since FIGS. 6(a), 6(b), and 6(c) correspond to FIGS. 3(a), 3(c), and 3(d), respectively, the above description will be referred to. In addition, since FIGS. 7(a), (b) and (c) correspond to FIGS. 3(a), (c) and (d), respectively, the above description will be referred to.

First, the resonance type power converter according to the reference example will be described. The resonance type power converter according to the reference embodiment has the same configuration and functions as the resonance type power converter according to the present embodiment except that the shunt circuit 3 is not provided and the controller 14 executes only the first switching element S₁. For convenience of explanation, the functions of the controller 14 are the same as those of the controller 4 shown in FIG. 1. In other words, the resonance type power converter according to the reference example is a so-called common class-E circuit.

As shown in FIG. 6, during turn-off period of the first switching element S₁, a voltage across both terminals of the first switching S₁ varies with time along a sine wave due to the charging and discharging of the shunt capacitor C_(s). The voltage V_(ds) applied across both terminals of the first switching element S₁ is shown in Equation (4) below.

$\begin{matrix} {\left\lbrack {{Equation}\mspace{14mu} 4} \right\rbrack \mspace{616mu}} & \; \\ {V_{ds} = \frac{\int{I_{cs}dt}}{C_{s}}} & (4) \end{matrix}$

However, V_(ds) is the voltage applied across both terminals of the first switching element S₁, I_(cs) is the current flowing into the shunt capacitor C_(s), and C_(s) is capacitance of the shunt capacitor C_(s).

Here, considering how to reduce the voltage applied across both terminals of the first switching element S₁, from Equation (4), it may be changed the current I_(cs) flowing into the shunt capacitor C_(s) or capacitance value of the shunt capacitor C_(s). However, since capacitance value of the shunt capacitor C_(s) is a parameter for executing soft switching operation, it is difficult to change capacitance value when trying to secure soft switching operation. Therefore, if an attempt is made to reduce the voltage applied across both terminals of the first switching element S₁, the current I_(cs) flowing into the shunt capacitor C_(s) is reduced. In the described embodiment, by controlling the second switching element S₂, it is possible to reduce the current I_(cs) flowing into the shunt capacitor C_(s), thereby reducing the voltages applied across both terminals of the first switching element S₁.

Next, the resonance type power converter according to the comparative example shown in FIG. 7 will be described. The resonance type power converter according to the comparative example is an apparatus for reducing the current flowing into the shunt capacitor, thereby reducing the voltage applied across both terminals of the first switching element. The resonance type power converter according to the comparative example comprises a resonance circuit that always superimposes harmonics of an integral multiple of the operating frequency of the first switching element. The resonance type power converter according to the comparative example is a so-called E/F class circuit. FIG. 7 (d) shows the current I′ flowing into the resonance circuit of the class E/F circuits.

As shown in FIG. 7, in the resonance type power converter according to the comparative embodiment, the current s flowing into the shunt capacitor is reduced by always shunting the current flowing into the shunt capacitor to the resonance circuit. As a result, the peak voltage applied across both terminals of the first switching element is reduced, while the rise speed of the voltage applied across both terminals of the first switching element is increased when the first switching element is turned off. In FIG. 7, the rising rate of the voltage applied across both terminals is indicated by the slope with respect to the voltage V_(ds) when the first switching element is turned off. The faster the rise speed of the voltage applied across both terminals increases the switching loss that occurs when the first switching element is turned off.

On the other hand, the resonance type power converter according to the present embodiment comprises the first switching element S₁ connected in parallel to the input voltage source 1, the series resonance circuit configured with the coil L₀ and capacitor C₀ (first resonance circuit) connected to the first switching element S₁, the shunt circuit 3 including the second switching element S₂ and shunting a current flowing into the shunt capacitor C_(s) in accordance with the operation of the second switching element S₂, the controller 4 for controlling the first switching element S₁ and the second switching element S₂. The controller 4 makes the current flowing into the shunt capacitor C_(s) to shunt by controlling the second switching element S₂ during a predetermined period within turn-off period of the first switching element S₁. This allows current to flow into the shunt capacitor C_(s) at the appropriate timing during turn-off period of the first switching element S₁, thereby reducing the switching losses that occur during soft switching operation.

FIG. 8 is an example of a comparison result of the resonance type power converter according to the present embodiment, the resonance type power converter according to the reference example described with reference to FIGS. 5 and 6, and the resonance type power converter according to the comparative example described with reference to FIG. 7. The energy loss includes a steady-state loss that occurs regularly in the first switching element S₁ and a switching loss that occurs when the first switching element S₁ is turned on and off. As shown in FIG. 8, in the present embodiment, as compared with the resonance type power converter according to the reference example and the resonance type power converter according to the comparative example, it is possible to significantly reduce the switching loss.

Also, in the present embodiment, the controller 4 turn on the second switching element S₂ later than the first switching element S₁ is turned off. This makes it possible to suppress rise speed of the voltage applied across both terminals of the first switching element S₁ when first switching element S₁ is turned off, and consequently to reduce the switching losses that occur when the first switching element S₁ is turned off.

In addition, in the present embodiment, the controller 4 turns off the second switching element S₂ earlier than the first switching element S₁ is turned on. This allows the current flowing into the shunt capacitor C_(s) to be shunted before the first switching element S₁ is turned on. As a result, the zero-voltage switching operation is ensured, and switching losses generated when the first switching element S₁ is turned on can be reduced.

In addition, in the present embodiment, the current flowing into the shunt capacitor C_(s) is shunted by utilizing the resonant phenomena in the series resonance circuit (the second resonance circuit) configured with the coil L and the capacitor C₁ included in the shunt circuit 3. Thus, the shunt circuit 3 can be configured by the resonance circuit configured with the coil L and the capacitor C₁, and the shunt circuit 3 can be miniaturized.

In the present embodiment, the controller 4 controls the operating frequency of the second switching element S₂ so that the resonance frequency of the second resonance circuit is higher than the operating frequency of the first switching element S₁. This allows the resonance frequency of the second resonance circuit to be set outside the operating band of the first switching element S₁ so as to restrain the effect on the zero-voltage switching operation of the first switching element S₁. In other words, the switching loss can be further reduced without changing the characteristics of the class E circuit.

Further, in the present embodiment, the controller 4 controls the second switching element S₂ in accordance with the inverse of the resonance frequency of the second resonance circuit. Thus, the turn-on period of the second switching element S₂ can be set to a real multiple of the period indicated by the reciprocal of resonance frequency of the second resonance circuit. In other words, the shunt can be realized by controlling the duty ratio of the turn-on period to the second switching element S₂. As a result, even when variations occur in the inductances of the coil L and capacitance values of the capacitor C₁ configured with the shunt circuit 3 due to changes in the ambient temperatures, for example, the currents flowing into the shunt capacitors C_(s) can be shunted with high accuracy.

The second resonance circuit according to the present embodiment is similar to the snubber circuit for restraining noises to the first switching element S₁, but since resonance frequency occurred by the second resonance circuit is set outside the operating band of the first switching element S₁, the operational advantage differs from the operational advantage by the snubber circuit.

Second Embodiment

Next, the resonance type power converter according to a second embodiment will be described. FIG. 9 is a diagram illustrating a configuration of a power conversion system comprising the resonance type power converter according to the second embodiment. The resonance type power converter according to the second embodiment has the same configuration and functions as the resonance type power converter according to the first embodiment except that the circuit configuration of the shunt circuit 5 and method for controlling the second switching element S₂ by the controller 6. For this reason, the descriptions used in the above-described embodiments will be referred to for configurations and functions like those of the resonance type power converter according to the first embodiment.

A shunt circuit 5 is connected in parallel to the shunt capacitor C_(s) to shunt the current flowing into the shunt capacitor C_(s). In this embodiment, the shunt circuit 5 is configured with a capacitor C₂ and a second switching element S₂ connected in serial to the capacitor C₂. One end of the capacitor C₂ is connected to the other end of the input coil L_(c) and the other end of the capacitor C₂ is connected to a source terminal of the second switching element S₂. A drain terminal of the second switching element S₂ is connected to a low-potential side output terminal of the input voltage source 1.

Capacitance of the capacitor C₂ is set to a value greater than capacitance value of the shunt capacitor C_(s). Thus, the current flowing into the shunt capacitor C_(s) can be flowed into the capacitor C₂. The current flowing into the capacitor C₂ will be described later.

The second switching element S₂ has a diode inside the device. As shown in FIG. 9, an anode terminal of the diode is connected to the source terminal of the second switching element S₂, and a cathode terminal of the diode is connected to the drain terminal of the second switching element S₂. In other words, unlike the first embodiment, the anode terminal of the diode is disposed on the high-potential side of the input voltage source 1, the cathode terminal of the diode is disposed on the low-potential side of the input voltage source 1. Thus, when the voltage V_(s1_ds) across both terminals of the first switching element S₁ is higher than a voltage V_(c2) across both terminals of the capacitor C₂, the current flowing into the shunt capacitor C_(s) can be automatically flowed through the diode into the capacitor C₂. Incidentally, the shunt circuit may have any circuit configuration that limits the voltage V_(s1_ds) across both terminals of the first switching element S₁ and is not limited to the circuit configuration shown in FIG. 9.

By controlling the second switching element S₂, the controller 6 makes the current flowing into the shunt capacitor C_(s) to shunt so as to cause the current to flow through shunt circuit 5.

The timing to turn on and off the second switching element S₂ and the turn-on period of the second switching element S₂ will be described with reference to FIG. 10.

FIG. 10 is a diagram for explaining a control signal to the second switching element S₂. Since FIG. 10 corresponds to FIG. 2, the description of each reference numeral is appropriately used.

As shown in FIG. 10, the controller 6 controls the second switching element S₂ while a control signal to the first switching element S₁ is low-level. The controller 6 turns on the second switching element S₂ for a predetermined period within turn-off period of the first switching element S₁ and turns off the second switching element S₂ after a predetermined time. Specifically, the controller 6 turns on the second switching element S₂ later than the first switching element S₁ is turned off. When the second switching element S₂ is on, the controller 6 turns off the second switching element S₂ earlier than the first switching element S₁ is turned on.

Also, in the present embodiment, the controller 6 turns on the second switching element S₂ in accordance with the relationship between the voltage V_(s1_ds) across both terminals of the first switching element S₁ and the voltage V_(c2) across both terminals of the capacitor C₂. Specifically, the controller 6 turns on the second switching element S₂ when the voltage V_(s1_ds) across both terminals of the first switching element S₁ is higher than the voltage V_(c2) across both terminals of the capacitor C₂. As a configuration capable of detecting each voltage, a configuration in which a voltage sensor (not shown) for detecting the voltage V_(s1_ds) across both terminals of the first switching element S₁ and a voltage sensor (not shown) for detecting the voltage V_(c2) across both terminals of the capacitor C₂ are provided and each voltage is input to the controller 6 can be exemplified.

In the example of FIG. 10, the controller 6 turns off the first switching element S₁ (time T₁), and when the voltage V_(s1_ds) across both terminals of the first switching S₁ is higher than the voltage V_(c2) across terminals of the capacitor C₂, the controller 6 turns on the second switching element S₂ (time T_(s2_ons)). The controller 6 then turns on the second switching element S₂ during the turn-on period T_(s2_on), and then turns off the second switching element S₂ (time T_(s2_one)). When the control to the second switching element S₂ is completed, the controller 6 turns on the first switching element S₁ (time T₂).

Next, with reference to FIGS. 9 and 11, the operation of the resonance type power converter according to the present embodiment. FIG. 11 is an exemplary operation of the resonance type power converter according to the present embodiment. FIGS. 11(a), (b), (d), and (e) correspond to FIGS. 3(a), (b), (d), and (e), and therefore, the description used in the above embodiment will be referred to. FIG. 11(c) shows the voltage V_(s1_ds) applied across the drain terminal and the source terminal of the first switching element S₁ and the voltage V_(c2) across both terminals of the capacitor C₂. The operation of the resonance type power converter is described in the order of the time series shown in FIG. 11.

The operation of the resonance type power converter from the turn-off of the first switching element S₁ to the turn-on of the second switching element S₂ will be described. As shown in FIGS. 11(b), (d), (e), when the first switching element S₁ is turned off, charging to the shunt capacitor C_(s) begins, and the voltages applied across both terminals of the first switching element S₁ connected in parallel to the shunt capacitor C_(s) begins to rise.

Next, the operation of the resonance type power converter when the second switching element S₂ is turned on will be described. As shown in FIGS. 11(b), (d), (e), when the second switching element S₂ is turned on by the controller 6 with the first switching element S₁ is off, the current I_(c) rises from the zero current and the current ICs becomes the zero current. This indicates that, as shown in FIG. 9, when the second switching element S₂ is turned on, the current flowing into the shunt capacitor C_(s) begins to flow into the capacitor C₂. The voltage V_(s1_ds) is then limited by the voltage values of the voltage V_(c2) applied across both terminals of the capacitor C₂.

Next, the operation of the resonance type power converter during the turn-on period of the second switching element S₂ will be described. As shown in FIG. 11(c), the voltage V_(s1_ds) is maintained by the voltage values of the voltage V_(c2). This indicates that during the turn-on period of the second switching element S₂, the current planned to flow into the shunt capacitor C_(s) is shunted, and the shunted current is flowing into a capacitor C₂ greater than capacitance value of the shunt capacitor C_(s). FIG. 3E shows that the capacitor C₂ is charged and discharged.

Incidentally, since the operation from when the second switching element S₂ is turned off to when the first switching element S₁ is turned on is the same as the operation in the first embodiment, the description of the first embodiment is appropriately incorporated.

As described above, in the resonance type power converter according to the present embodiment, by controlling the second switching element S₂ during turn-off period of the first switching element S₁, the current flowing into the shunt capacitor C_(s) can be shunted to limit the voltage applied across both terminals of the first switching element S₁. By reducing the peak voltage value, it is possible to reduce the load on the first switching element S₁.

Next, with reference to FIG. 12, the input impedance characteristics of the resonance type power converter according to the present embodiment will be described. FIG. 12 is an input impedance characteristic of the resonance type power converter according to the present embodiment. FIG. 12 is the input impedance characteristic when viewing the load 2 side from the input voltage source 1 among the resonance type power converter shown in FIG. 9.

The operating band of the first switching element S₁ shown in FIG. 12 is a frequency band for executing a soft switching operation. In this embodiment, capacitance value of the capacitor C₂ is set greater than capacitance value of the shunt capacitor C_(s). Thus, the impedance characteristics of capacitance of the capacitor C₂ are higher frequency bands than the operating band of the first switching element S₁. In FIG. 12, the sub-resonance point defined by the capacitor C₂ is not shown.

As described above, in the present embodiment, by accumulating charges in the capacitor C₂ included in shunt circuit 5, the current flowing into the shunt capacitor C_(s) is shunted. Thus, it is possible to reduce the number of elements configured with the shunt circuit 5, as a result, the cost of the shunt circuit 5 can be reduced.

Also, in the present embodiment, the voltage applied across both terminals of the first switching element S₁ is limited by using a capacitor with a capacitance value greater than capacitance value of the shunt capacitor C_(s) as the capacitor C₂. This allows the impedance property of capacitance of the capacitor C₂ to be set outside the operating band of the first switching element S₁, thereby restrain the effect on the zero-voltage switching operation of the first switching element S₁. In other words, the switching loss can be further reduced without changing the characteristics of the class E circuit.

In addition, in the present embodiment, the controller 6 turns on the second switching element S₂ when the voltage V_(s1_ds) across both terminals of the first switching element S₁ is greater than the voltage V_(c2) across both terminals of the capacitor C₂. Thus, since the capacitor C₂ having a large capacitance is charged, the electric charge stored in the shunt capacitor C_(s) can be extracted. As a result, it is possible to restrain a voltage rise across both terminals of the shunt capacitor C_(s) and a voltage rise across both terminals of the first switching element S₁.

It should be appreciated that the embodiments explained heretofore are described to facilitate understanding of the present invention and are not described to limit the present invention. It is therefore intended that the elements disclosed in the above embodiments include all design changes and equivalents to fall within the technical scope of the present invention.

For example, in the embodiment described above, the configuration using the input voltage source 1 has been described as an example, but the present invention is not limited to this. For example, an input current source may be used instead of the input voltage source 1 and the input-coil L_(c).

For example, in the first embodiment described above, the turn-on period T_(s2_on) of the second switching element S₂ has been described with reference to a configuration that is set to satisfy the equations (1) to (3), but the present invention is not limited to this. The turn-on period T_(s2_on) may be set to satisfy only Equation (1).

For example, in this specification, the resonance type power converter according to the present invention has been described exemplifying the inverter, but the present invention is not limited to this. In this specification, the first switching element according to the present invention has been described by exemplifying the first switching element S₁, but the present invention is not limited to this. In this specification, the second switching element according to the present invention has been described by exemplifying the second switching element S₂, but the present invention is not limited to this. In this specification, the first resonance circuit according to the present invention has been described by exemplifying the series resonance circuit configured with the coil L₀ and the capacitor C₀, but the present invention is not limited to this. In this specification, the second resonance circuit according to the present invention has been described by exemplifying the series resonance circuit configured with the coil L and the capacitor C₁, but the present invention is not limited to this. In this specification, the shunt circuit according to the present invention has been described by exemplifying the shunt circuit 3 and 5, but the present invention is not limited to this. In this specification, the first capacitor according to the present invention has been described by exemplifying the shunt capacitor C_(s), but the present invention is not limited to this. In this specification, the second capacitor according to the present invention has been described by exemplifying the capacitor C₂, but the present invention is not limited to this.

EXPLANATIONS OF LETTERS OR NUMERALS

-   10 . . . Power conversion system -   1 . . . Input voltage source -   2 . . . Load -   3 . . . Shunt circuit, -   4 . . . Controller 

1.-10. (canceled)
 11. A method for controlling a resonance type power converter that includes a first resonance circuit and a shunt circuit, converts and outputs power of a DC power supply, the method for controlling the resonance type power converter comprising: shunting a current flowing into a first capacitor by controlling a second switching element during a predetermined period within a turn-off period of the first switching element, the first capacitor connected in parallel to the first switching element, the second switching element included in the shunt circuit, and the first switching element operated in response to resonance of the first resonance circuit.
 12. The method for controlling the resonance type power converter according to claim 11, comprising: turning on the second switching element later than the first switching element is turned off.
 13. The method for controlling the resonance type power converter according to claim 11, comprising: turning off the second switching element earlier than the first switching element is turned on.
 14. The method for controlling the resonance type power converter according to claim 11, comprising: shunting the current by resonating a second resonance circuit included in the shunt circuit.
 15. The method for controlling the resonance type power converter according to claim 14, comprising: controlling the second switching element such that a resonance frequency of the second resonance circuit is higher than a switching frequency of the first switching element.
 16. The method for controlling the resonance type power converter according to claim 14, comprising: setting the predetermined period for controlling the second switching element in accordance with an inverse of a resonance frequency of the second resonance circuit.
 17. The method for controlling the resonance type power converter according to claim 11, comprising: shunting the current by storing electric charge in a second capacitor included in the shunt circuit.
 18. The method of controlling the resonance type power converter according to claim 17, comprising: limiting a voltage across both terminals of the first switching element using a capacitance of the second capacitor that is greater than a capacitance of the first capacitor.
 19. The method for controlling the resonance type power converter according to claim 17, comprising: turning on the second switching element when a voltage across both terminals of the second capacitor is higher than a voltage across both terminals of the first switching element.
 20. A resonance type power converter comprising: a first switching element connected in parallel to a DC power supply; a resonance circuit connected to the first switching element; a first capacitor connected in parallel to the first switching element; a shunt circuit including a second switching element and shunting a current flowing into the first capacitor in response to an operation of the second switching element; and a controller configured to control the second switching element, the controller being further configured to control the second switching element during a predetermined period within a turn-off period of the first switching element operating in response to resonance of the resonance circuit.
 21. The method for controlling the resonance type power converter according to claim 12, comprising: turning off the second switching element earlier than the first switching element is turned on.
 22. The method for controlling the resonance type power converter according to claim 15, comprising: setting the predetermined period for controlling the second switching element in accordance with an inverse of a resonance frequency of the second resonance circuit.
 23. The method for controlling the resonance type power converter according to claim 18, comprising: turning on the second switching element when a voltage across both terminals of the second capacitor is higher than a voltage across both terminals of the first switching element. 